1. Field of the Invention
The invention relates to active common mode EMI filter circuits using bipolar transistors or MOSFETs in combination with common mode current sensing transformers.
2. Background Art
Various circuits of this type, having a unity gain amplifier in a feedforward configuration, or a high-gain amplifier in a feedback configuration, have previously been disclosed.
Active EMI filters are known and are described, for example, in co-pending applications Ser. No. 09/816,590, filed Mar. 23, 2001 entitled ACTIVE FILTER FOR REDUCTION OF COMMON MODE CURRENT (IR-1744), and Ser. No. 10/336,157, filed Jan. 2, 2003, entitled ACTIVE EMI FILTER WITH FEED FORWARD CANCELLATION (IR-2146), the disclosures of which are incorporated herein by reference.
Active EMI filter circuits may use the feed forward design as disclosed in the above noted application Ser. No. 09/816,590. See FIGS. 1A and 1B. The feed forward design, shown in FIG. 1B, has fundamentally superior performance characteristics over the traditional feedback design, which is shown in FIG. 1A.
FIG. 1A shows the prior art feedback configuration. In the feedback configuration of the active EMI noise filter for reducing common mode current, the noise sensor may comprise, for example, a current transformer CT which has two primaries each coupled to a respective leg of the DC bus fed rectified DC current by a rectifier circuit R from the AC mains. Each primary is coupled in series with the respective DC bus. Each DC bus is coupled to an inverter I which is controlled to provide three phase AC current to a load, e.g., an electric motor M.
The secondary winding of the current transformer CT is coupled to an amplifier A. The common mode current is the noise current that flows in common in the same direction in both legs of the DC bus to the inverter and is caused by intrinsic reactive components (typically capacitive) between the motor windings and the motor case and/or inverter heat sink. Without filtering via capacitor CFFILT, this common mode current will be returned to the mains network ground GND and reflected as undesirable noise currents on the AC mains. The aim of the active EMI filter circuit is to provide a path for the common mode current via capacitor CFILT though an active switching circuit back to the DC bus and hence contain the common mode current as a circulating current in the DC bus, inverter and motor and prevent its conduction back to the AC network. The common mode current returned to ground GND is thus canceled.
The amplifier A outputs of FIG. 1A control two complementary transistors Q1 and Q2 to shunt common mode current through the capacitor CFILT from the ground line L connecting the motor case and ground. The common mode current to be minimized comprises a commonly polarized current which flows in each leg of the DC bus through the inverter to the motor M and is attributable to current flows between the windings of the motor and the motor case/inverter heat sink due to the internal intrinsic capacitances. The common mode current then flows back through the ground line and would ordinarily flow to ground. This would create unnecessary noise currents and harmonics on the AC lines. In order to minimize these currents, passive filters have been used in the past to shunt the common mode current. FIG. 1A shows a prior art active feedback filter arrangement in which the common mode current is shunted by the capacitor CFILT through either transistor Q1 or Q2, depending upon which transistor is turned on (which depends on the common mode current flow direction at a particular instant), back to the DC bus, thereby eliminating the common mode current returned to ground, and thereby minimizing noise currents reflected in the AC lines. However, the system shown in FIG. 1A requires the amplifier A and current transformer to have high gain in order to minimize the common mode current returned to ground. Theoretically, the gain of the system is required to be infinite to have the common mode current returned to ground equal to zero, as shown by the equations in FIG. 1A and the equivalent circuit of FIG. 1Aa. This results in possible oscillation of the system and furthermore requires a current transformer of moderate size. Furthermore, the signal to noise ratio is low, consistent with the need for the high gain amplifier.
In contrast, with the feed forward arrangement as shown in FIG. 1B, and the equivalent circuit of FIG. 1Ba, the common mode current returned to ground GND from line L is theoretically zero for a system gain equal to 1. Accordingly, the amplifier gain is easy to attain, the system has excellent stability and because of the high signal to noise ratio, a relatively small current transformer can be utilized. Accordingly, the feed forward design shown in FIG. 1B has significant advantages because the current transformer can be of significantly smaller size, the amplifier is easier to embody and the system has good stability and is free from oscillation.
Another example of a previously disclosed circuit, which uses two N-Channel MOSFETs, and a current-sensing transformer with 1-turn primaries and secondaries, is shown in FIG. 2A. This circuit is potentially attractive over circuits that use bipolar transistors or P-channel MOSFETs, because N-channel MOSFETs are available with voltage ratings that are higher than those of bipolar transistors or P-channel MOSFETs. The MOSFET's greater ruggedness versus bipolar transistors is also advantageous. For even higher voltage, IGBTs might be preferred over MOSFETs, because of their higher transconductance and higher peak current capability.
A problem with MOSFETs or IGBTs, however, is that the gate to source drive voltage needed to deliver the required output current is relatively high. For example, the peak gate-source voltage required to drive a 200V IRFD210 HEXFET to an output current of 2.5 A peak is about 6.5V. A 500V IRF820 would require about the same gate-source voltage for 2.5 A peak output current.
The gate-source voltage for the MOSFET is induced across the secondary windings S1 (for Q1) and S2 (Q2) of the current sensing transformer. For this unity current gain feedforward configuration the primary to secondary turns ratio must, by definition, be 1.0. The voltage induced across the primary winding is thus the same as the secondary voltage; in the above examples, this voltage would be about 6.5V.
Unfortunately, the higher the voltage across the primary, the greater the magnetizing component of the primary current. Since the current that flows in the secondary is equal to the total primary current less the magnetizing component, the greater the voltage across the primary, the greater the error between the secondary current—which is the output current of the active filter—and the primary current. The greater the error between the output current of the active filter and the primary common mode current, the poorer the performance of the active filter.
FIG. 2B shows waveforms of iCOMDRIVE and the output current, iOUT of the active filter, with one primary and one secondary turn, on a Magnetics ZW-42507 toroid ferrite core. A significant error exists between iOUT and ICOMDRIVE.
A way of minimizing the error due to the magnetizing current, shown in FIG. 2C, is to increase the number of turns on the current sensing transformer—keeping a 1:1 ratio between the primary and secondary turns, in order to preserve unity current gain. In this example, each winding has 3 turns. Since the voltage across the winding is set by the gate-source voltage of the MOSFET, and therefore is fixed regardless of the number of turns, the magnetizing component of current decreases in inverse proportion to the square of the number of turns (because the magnetizing inductance is proportional to the square of the turns).
FIG. 2D shows corresponding waveforms with three primary and three secondary turns. The error between iOUT and iCOMDRIVE has been substantially reduced. The disadvantage is that a multiple turn hand-wound primary is now needed.
It is desirable, however, for the number of primary turns to be minimized. Optimally, just a single primary wire, running through the center of a toroid (i.e., a single ‘turn’) is desired. The reason is that the primary wire must have relatively large diameter, because it must be rated to carry the full normal-mode current. A multiple turn primary of large diameter wire has to be hand-wound onto the toroid, and this is expensive.
Another known configuration of an active common mode filter, using bipolar transistors, is shown in FIG. 3. Like the circuits in FIGS. 2A and 2C, this is a “feedforward” circuit, which has unity gain between iOUT and iCOMDRIVE.
It is desirable for the primary winding on the common mode sensing transformer to be a single wire that passes through the center of the magnetic core. The reason is that the primary winding has relatively large cross section, because it carries the full normal mode current of the drive. The secondary current of the CT is a signal that represents just the common mode current, and has low average value. Thus, the secondary wire can have much smaller cross-section than the primary.
Since the current gain must be unity, a single primary ‘turn’ on the current sensing transformer requires that the secondary also have just one ‘turn’. Unfortunately, a single-turn current-sensing transformer requires a magnetic core with an inconveniently large cross-section. The reason is that the voltage developed across the secondary, hence also the primary voltage, is essentially the base-emitter voltage of the transistors Q1/Q2 (because the turns ratio is 1.0). This voltage is typically about 1V. This voltage developed across the one-turn primary creates a significant component of magnetizing current, because the magnetizing inductance of one turn is relatively low—unless the cross section of the core is made inconveniently large.
The magnetizing component of common mode current in the primary is not transmitted to the secondary, thus it becomes an “error” in the output current iOUT of the amplifier. This degrades the performance of the active filter.
In order to use a small inexpensive common mode current sensing transformer with just one primary turn, it is necessary to find a way of reducing the voltage developed across the primary, thus of reducing the magnetizing current.
One approach is to create an offsetting bias voltage in the base-emitter circuit of the transistors, which cancels the base-emitter threshold voltage, as described below in connection with FIG. 4(a). This approach accurately matches the bias voltage to the base-emitter voltage of the transistors over the required range of operating temperature.
Another approach (not shown in the drawings) is to use multiple turns on the secondary of a one-turn primary sensing transformer. The voltage reflected back to the primary winding is now Vb-e/N, where N is the number of secondary turns. The primary component of magnetizing current, relative to the total common mode primary current, is now reduced by 1/N. The ratio of error to signal of the secondary current is thus also reduced by       1    N    ,though the absolute amplitude of the secondary current is also reduced by   1  Ni.e. the secondary current is a relatively more accurate, yet miniaturized, replica of the primary current.
However, since the overall current gain from iCOMDRIVE to iOUT must be unity, some form of current amplifier, such as a current mirror circuit, designed to have current gain of N, is required to restore equality between iOUT and iCOMDRIVE. A problem with this type of current mirror is that it has inaccuracies due to tolerances in the matching of impedances and/or transistor characteristics.